In bandwidth efficient digital communication systems the effect of each symbol transmitted over a time dispersive channel extends beyond the time interval used to represent that symbol. The distortion caused by the resulting overlap of received symbols is called intersymbol interference (ISI). Intersymbol interference (ISI) arises in all pulse modulation systems, including frequency shift keying (FSK), phase shift keying (PSK) and quadrature amplitude modulation (QAM) and pulse amplitude modulation (PAM).
Band limiting of the transmission channel results in the data signal at the receiver being subject to distortion caused by intersymbol interference (ISI).
When transmitting data pulses via a data transmission channel severe signal distortion caused by band limiting of the transmission channel, signal reflections (echo) are caused by incorrect matching to the line impedance both at the transmitting end and at the receiving end of the data transmission channel and resonances in the transmission characteristic of the data transmission channel degrade the received signal.
The purpose of an equalizer placed in the path of the received signal is to reduce the intersymbol interference (ISI) as much as possible to maximize the probability of correct decisions. Accordingly an equalizer is a filter to reduce the effect of intersymbol interference.
There are many different type of equalizers. The simplest form of an equalizer is a linear transversal equalizer wherein the current and passed values of the received data signal are linearly weighted by equalizer coefficients which are summed up to produce an equalizer output signal.
Non linear decision feedback equalizers (DFE) are particularly useful for data transmission channels which severe amplitude distortion. A decision feedback equalizers (DFE) uses decision feedback to cancel the interference from symbols which have been already detected. The decision feedback equalizer comprises a forward circuit part and a feedback circuit part and the equalized signal is formed by the sum of the outputs of the forward and feedback parts. The forward circuit part of the decision feedback equalizer is formed like a linear transversal equalizer as discussed above. Decisions made on the equalized signal are fed back via a second transversal filter. The basic idea of decision feedback equalizers is that if the values of the symbols are already detected are known (past decisions assumed to be correct) then the intersymbol interference contributed by these symbols can be canceled exactly, by subtracting path symbol values with appropriate weighting from the equalizer output.
In contrast to lower-frequency applications for data transmission (for example 2 Mbit/s band), complex equalization of the data stream upstream of a data recovery unit (for example clock and data recovery, CDR) is not possible. Conventional methods which are based on over-sampling by means of analog/digital converters and digital equalization (for example Viterbi decoders, Decision Feedback Eq. Etc.) can generally not be used for very high data rates, for technological reasons (maximum bandwidth of semiconductor technology).
At present, the preamplifiers for data transmission applications via PBC or back-plane connections are generally simple limiting amplifiers. They are provided to maximize the eye opening, that is to say the amplitude of the signal, in order that the downstream data recovery unit has as much signal energy as possible with the best possible signal-to-noise ratio (SNR) for evaluation at the decision time.
Increasing the SNR by signal limiting is valid only for as long as the input signal to be amplified is not subject to major edge an/or amplitude noise. However, in reality, the signal is generally noisy. The consequence of simple limiting is AM-to-PM conversion, that is to say conversion of the amplitude error to increased edge noise. Error-free recovery of the signal thus becomes considerably more difficult for the downstream data recovery unit, in this case generally a clock and data recovery circuit or timing recovery circuit. A worse bit error rate can generally be expected. The maximum possible range and the maximum permissible attenuation of the channel are in consequence restricted. This is contrary to the requirement for high data rates of more than 1 Gbit/sec over relatively highly band-limited transmission paths.
A proposed prior art solution is to use complex adaptation methods. There are two major options for this—a linear feed forward equalizer (FFE) or an equalizer with feedback (DFE), such as decision feedback.
Other conventional equalizers, which are based on statistical methods, can be used only to a limited extent at these high data rates owing to the lack of additional information in the data stream, for example a preamble. This is a purely stochastic data stream.
The implementation of the sated prior art adaptation methods—for example decision feedback (DFE) is considerable complex.
FIG. 1 shows a feed forward equalizer (FFE) according to the state of the art.
The transmitter sends a data signal via a data transmission channel to the input of the feed forward equalizer as shown in FIG. 1. The feed forward equalizer (FFE) is provided to suppress intersymbol interferences (ISI) caused by the data transmission channel. For this purpose the feed forward equalizer (FFE) comprises a low path filter (LPF) which is connected in series to a first amplifier A1 the output of which is connected to substracting means to substract the filtered and amplified signal from the received signal buffered by a second amplifying means A2. The equalized output signal of the feed forward equalizer (FFE) is output to a decision unit of a receiver. The low path filter (LPF) employed by the feed forward equalizer (FFE) according to the state of the art as shown in FIG. 1 can be an analog or passive low path filter.
The analog equalization method according to the present invention combines discrete time methods with continuous-time methods for equalization of the data signal, in particular for high-speed serial data transmission, in which quantization of the signal is impossible or is too complex. The method according to the invention offers a capability for simple signal equalization without feedback. In most (multichannel) high-speed links, the necessary phases for the equalizer are required in any case for the subsequent timing recovery. Only a small amount of additional circuit complexity is thus required. Furthermore, this unit can be designed such that it can be adjusted as required—until the circuit function is reduced to that of a normal limiting amplifier. In addition, the gain levels/weighting coefficients of the feed forward equalizer according to the invention can be programmed.
FIG. 2 shows the timing of a signal profile. A trapezoidal signal is applied to the data transmission channel at the transmitter. The band limiting in the transmission path leads to the impulse response as shown in FIG. 2b. The trailing edge of the impulse response leads, with a high data density, to superimpositions of the individual impulse responses—the signal Uin is distorted. The desired impulse response Uout provided to the receiver is illustrated at FIG. 2c. In order to obtain this impulse response UOUt, it is necessary to subtract the appropriate time components from the actual impulse response.
The feed forward equalizer (FFE) according to the state of the art as shown in FIG. 1 does not eliminate sufficiently the distortions caused by the data transmission channel for data signals with a high data rate (DR) of more than one gigabit per second (1 Gbit/sec).
FIG. 3a shows the poles of the signal path formed by the low path filter (LPF) and the amplifier A1 in the complex plane.
Beside the pole of the low path filter the band limitation of the operation amplifier A1 forms a parasitic pole. The ratio between the parasitic pole and the desired pole formed by the low path filter (LPF) should always be higher than ten
      (                            W          p                          W          par                    >      10        )    .Because of the band limitation of the operation amplifier A1 the parasitic pole approximates the pole of the low path filter (LPF) with an increasing data rate (DR). The basic problem of the feed forward equalizer (FFE) according to the state of the art is the band limitation of the analog elements.
FIG. 3b shows a bode diagram of the feed forward equalizer (FFE) according to the state of the art as shown in FIG. 1. As can be seen from FIG. 3b the parasitic pole caused by the operation amplifier A1 causes a steeper amplitude decrease in the higher frequency range. The phase shift of the FFE approximates −180° with increasing data rate DR. Accordingly the group delay time is no longer constant and the impulse response of the feed forward equalizer (FFE) becomes unsymmetric with the increasing data rate DR. The unsymmetric pulse response of the feed forward equalizer (FFE) significates that the intersymbol interferences (ISI) caused by the data transmission channel can no longer be equalized by the equalizer.
FIG. 4 illustrates the amplitude characteristic of the conventional feed forward equalizer (FFE) as shown in FIG. 1. When the data rate frequency of the transmitted data signal reaches the fundamental frequency of the operation amplifier the FFE equalizer according to the state of the art does not behave like an ideal equalizer, i.e. it does not compensate the attenuation of the data transmission channel by a reverse amplitude characteristic. The real forward equalizer (FFE) according to the state of the art comprises a −3 dB alternation at the fundamental frequency. When the parasitic pole of the amplifier approaches the fundamental frequency distortions occur.
When using conventional continuos-time equalization methods the equalizer comprises a transfer function which is inverse to the transfer function Hchannel of the data transmission channel. Mathematically, this can be described in the Laplace plane as follows:
                                          H            equ                    ⁡                      (            s            )                          =                              1                                          H                channel                            ⁡                              (                s                )                                              =                                    A              ·                                                                    ∏                    n                                                        k                    =                    1                                                  ⁢                                  (                                      1                    +                                          s                      /                                              ω                        zk                                                                              )                                                                                                      ∏                  m                                                  k                  =                  1                                            ⁢                              (                                  1                  +                                      s                    /                                          ω                      pk                                                                      )                                                                        (                  Equation          ⁢                                          ⁢          1                )            
The following transfer function is obtained for the feed forward equalizer (FFE) shown in FIG. 1.
                                          H            equ                    ⁡                      (            s            )                          =                                                            U                out                                            U                                  i                  ⁢                                                                          ⁢                  n                                                      ⁢                          (              s              )                                =                                    A              2                        -                                          A                1                            ⁢                              1                                  (                                      1                    +                                          s                      /                                              ω                        lk                                                                              )                                                                                        (                  Equation          ⁢                                          ⁢          2                )                                                      H            equ                    ⁡                      (            s            )                          =                                                            U                out                                            U                                  i                  ⁢                                                                          ⁢                  n                                                      ⁢                          (              s              )                                =                                    A              2                        -                                          A                1                            ⁢                              1                                                                            ∏                      m                                                              k                      =                      1                                                        ⁢                                      (                                          1                      +                                              s                        /                                                  ω                          lk                                                                                      )                                                                                                          (                  Equation          ⁢                                          ⁢          3                )            
Equation (3) illustrates the general representation of the implementation variant of the transfer function Hequ for approximation of the channel transfer function Hchannel. This implementation has the advantage that pole positions are required only for approximation—this is particularly advantageous for stability criteria.
In a discrete-time representation, the transfer function according to equation (3) can be represented as follows:
                                          H            equ                    ⁡                      (            z            )                          =                                                            U                out                                            U                                  i                  ⁢                                                                          ⁢                  n                                                      ⁢                          (              z              )                                =                                                    A                2                            -                                                A                  1                                ⁢                                  1                                      1                    +                                                                                            ∏                          n                                                                          k                          =                          1                                                                    ⁢                                              (                                                                              b                            bk                                                    ·                                                      z                                                          -                              kr                                                                                                      )                                                                                                                  =                                          A                2                            -                                                A                  1                                ⁢                                  1                                      1                    +                                                                  b                        b1                                            ·                                              z                                                                              -                            1                                                    ⁢                          τ                                                                                      +                                                                  b                        b2                                            ·                                              z                                                                              -                            2                                                    ⁢                          τ                                                                                      +                                          …                      ⁢                                                                                          ⁢                                                                        b                          bn                                                ·                                                  z                                                                                    -                              n                                                        ⁢                                                                                                                  ⁢                            τ                                                                                                                                                                                                      (                  Equation          ⁢                                          ⁢          4                )            
Equation (4) provides the basic formal relationship to implement the feed forward equalizer according to the present invention.
As has been shown above a conventional feed forward equalizer do not provide sufficient equalization for very high data rates of more than one gigabit per second.